Power electronic handbook

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20 Gate Drive Circuitry for Power Converters 551

15V

PWM2

PWM1

D2 R3

Dz4

Dz3

Dz2

Dz1

DC Bus

IGBT1

IGBT2

LOAD

D1 R1

FIGURE 20.14 Transformer coupled gate driver used to supply both the control signal and the gate drive power to the device.

15V

PWM

DC Bus

IGBT1

LOAD

Dz1

Dz2

FIGURE 20.15 Transformer coupled gate driver with a large duty cycle operating range.

purpose as described in Fig. 20.12. The second function

of this passive resistor network, is to dampen ringing

effects. This is caused by the interaction between the

IGBT or power MOSFET gate capacitance, and the gate

drive transformer leakage inductance.

• Transformer Coupled Gate Driver with Large Duty

Cycle Capability

Transformers offer excellent noise immunity and pro-

vide simple and cost-effective gate drive solutions, whilst

maintaining electrical isolation between the control and

gate drive electronics. A drawback however is the limita-

tion the transformer places on the maximum operating

duty cycle. Figure 20.15 offers a simple but effective solu-

tion to the conventional limitations by the introduction

of a DC restorer circuit which is formed by C2, Dz1, and

Dz2. This system allows for removal of any DC infor-

mation via C1, and restores the input waveform applied

with the addition of a negative voltage bias needed for

the IGBT gate drive. A small ferrite transformer core

can be used for a MOSFET gate driver operational to

several hundred kilohertz. This circuit can be redesigned

for bridge topologies, but is also well suited for high volt-

age DC–DC converters requiring a high-side switch. The

effective duty cycle range of this driver is from 5 to 95%.

Operating waveforms are shown in Fig. 20.16.

It should be noted that the gate drive voltage is

clamped at ﬁxed levels regardless of the duty cycle used,

unlike the case in Fig. 20.9. This technique also supplies

both the level shifted signal as well as the gate drive power,

eliminating the need for an additional ﬂoating supply.

The transformer turns ratio (T1) can also be adjusted to

allow the circuit in Fig. 20.15 to operate from a 5 V sup-

ply, whilst generating an output voltage swing from +15

to −5 V at the IGBT gate.

•

Transformer Coupled Signal Modulated Gate Driver

The circuit in Fig. 20.17 employs a high-frequency car-

rier signal that is modulated by a lower frequency

control signal (PWM). This is used to generate on/off

switching instants of power IGBT1. By employing a high-

frequency carrier, the transformer size is reduced, and

552 I. Khan

–V2

(a) 5% duty cycle (b) 50% duty cycle (c) 95% duty cycle

PWM

Vgate

FIGURE 20.16 Operating waveforms of the transformer coupled gate driver in Fig. 20.15.

D4D2

Vsec

Vout

Vgate

IGBT1

LOAD

DC Bus

U1B

U1A

PWM

VSQ1

FIGURE 20.17 Signal modulated carrier used for level shifting and generation of a ﬂoating supply.

Sec

SQ1

PWM

Out

Gate

Vtrigger

FIGURE 20.18 Signal modulated carrier used for level shifting and generation of a ﬂoating supply.

by modulating the time in which the carrier operates,

it controls the energy delivered to the gate of the IGBT.

A carrier frequency for VSQ1 should be chosen to be

much higher than the frequency of the PWM control

signal. When the PWM control signal is enabled, the

carrier signal is transformed to the secondary of trans-

former T1, which is rectiﬁed and ﬁltered to produce a

DC signal V

out

When the PWM control signal goes from the on- to

the off-state, the charge stored in the ﬁlter capacitor C1

discharges by a time constant determined by R1. This

is sometimes problematic when fast switching times

are required, especially in inverter bridge conﬁgura-

tions. A solution to this problem is to employ an

active driver (U1) on the secondary side of the trans-

former. This will detect the carrier, and switch the IGBT

20 Gate Drive Circuitry for Power Converters 553

20 k

U1B

U1A

+15V

Gate

Drive

cct

100

12 k

FIGURE 20.19 Low cost high frequency switch-mode ﬂoating supply.

gate accordingly. Operating waveforms for this circuit is

shown in Fig. 20.18 [1].

• High Frequency Floating Power Supply

Often gate driver systems require extra electronics which

has to be referenced to the ﬂoating switch being driven.

Extra functional electronics often leads to higher power

consumption, resulting in the need for a small low cost

ﬂoating power supply as shown Fig. 20.19. The input

section consisting of U1B, forms an oscillator used to

drive a MOSFET at high frequency. This MOSFET drives

a high-frequency transformer, which forms the isolation

medium between the common auxiliary power supply

and the ﬂoating secondary circuit. Transformer T1 pro-

duces a secondary output voltage which is rectiﬁed to

form a ﬂoating DC supply for the associated gate drive

circuitry. These ﬂoating power supplies are also avail-

able as monolithic DC–DC converter ICs with isolated

outputs.

20.4.2 Electronic Gate Drivers

These drivers utilize electronics in order to store energy in

capacitors. This is used to produce ﬂoating referenced poten-

tials. All circuits are referenced to a common ground potential.

This technique provides a very cost-effective gate driver solu-

tion over the isolated versions, and are becoming increasingly

popular in industries.

•

Gate Driver with Bootstrapped Floating Supply

The addition of a bootstrapped capacitor voltage allows

for the generation of a ﬂoating supply in Fig. 20.20.

Charging of C1 is achieved in the direction indicated

during the period when MOSFET M2 is turned on. The

turn-off of M2, results in the entrapment of charge in

C1 which now acts as the power supply to OPTO1, U1,

and the MOSFET M1’s gate. As current is drawn from

C1, the capacitor voltage tends to drop and needs to

be replenished cyclically. The selection of the bootstrap

capacitor value is critical for reliable operation of this

circuit under extended duty cycle conditions. Diode D1

sees the full DC bus voltage when M2 is off, and therefore

has to have a sufﬁciently high breakdown voltage. Appli-

cations with fast switching speed, require that D1 be a

fast recovery diode in order to withstand the high dv/dt

present across M2. Level shifting of the switching control

signal is achieved by means of the opto-couplers in the

circuit, providing the necessary control circuit isolation

required.

• Gate Driver with Floating Supply Derived from DC Bus

When the generation of a ﬂoating supply is required with-

out the need for isolation, the gate drive power can be

derived from the DC bus voltage. The circuit is charged

during the period when IGBT1 is off, with the gate drive

energy stored in C2. The circuit connected to the enable

pin of the opto-coupler, forms an under-voltage lock-

out which inhibits the IGBT drive signal at low DC

supply voltages, typically during the initial start-up of

the inverter circuit. The control signals are level shifted

through an opto-coupler which drives a totem pole

complimentary buffer stage as shown in Fig. 20.21 [8].

20.5 Current Technologies

The evolution of power semiconductor technology has led to

the development of smart driver IC modules. These modules

offer an essentially single chip/package solution to the gate

driver designer, with the inclusion of minimal external compo-

nents. These smart driver ICs utilize sophisticated electronics,

yielding an intelligent gate driver in a small module. Some of

these functions include the electronics for generating the ﬂoat-

ing supply from the auxiliary power supplied to the IC, level

shifting with or without isolation. Advanced on-board inte-

grated functions include under-voltage lockout, thermal trip,

overload and de-saturation detection, soft start, and shutdown.

These features enhance the effective utilization of the power

converter and also increase its reliability.

554 I. Khan

−

DC Bus

15Vdc

PWM1

PWM2

LOAD

OPTO1

OPTO2

FIGURE 20.20 Optically isolated level shifter with an un-isolated bootstrapped ﬂoating supply derived from the low-side power supply.

OUT

VCC

VEE

VEEB

HCPL2200

15V

R14

R13

Load

IGBT1

IGBT2

R10

R11

LED2

PWM2

PWM1

BJT1

BJT2

R12

LED1

Dz1

15V

DC Bus

FIGURE 20.21 Level shifted gate driver with ﬂoating supply derived from the DC bus.

20.5.1 Transformer Coupled Isolated Drivers

Semikron Corporation has developed a gate driver module

that employs integrated ﬂoating supplies (Fig. 20.22) [9]. The

level shifting is accomplished through transformer coupling,

which is used as a bi-directional communications link for the

transmission of the gate drive and fault signals.

Each module is conﬁgured for a half-bridge inverter con-

ﬁguration and one additional driver module can be added

for each inverter leg needed. The modules come in a variety

of output current capabilities, depending on the IGBT drive

requirements, with usable operating frequencies ranging from

the low kHz range up to 30 kHz.

20 Gate Drive Circuitry for Power Converters 555

input1

(TOP)

input2

(BOTTOM)

-input buffer

-short pulse

supression

-pulse shaper

-Vs monitor

-Error monitor

-Error memory

-interlock

-deadtime

SELECT

TDT1

TDT2

Error

input

When SKH122B is driving 1700V IGBTs, a 1kW/0,4W RVCE-resistor must be connected in series to the VCE-input.

The VCE terminal is to be connected to the IGBT collector C. If the VCE-monitoring is not used, connect S1 to S9 or S20 to S12 respectively.

Terminals P5 and P6 are not existing for SKH122A/21A; internal pull-up resistor exists in SKH122A/21A only.

Connections to SEMITRANS GB-models

integrated in ASIC

VCE

6k8/100

P12

3k2

GND/OV

S14

S13

S12

off

SEMITRANS

IGBT-Module

S15

S20

P14

P10

P13

Isolation

over

current

Power

driver

Power

driver

over

current

primary side secondary side output

LOAD

output2

(BOTTOM)

output1

(TOP)

ERROR

FIGURE 20.22 SKHI22 integrated half-bridge IGBT module capable of operation at 1200 V. A product of the Semikron Corporation.

Voltage

Supply

Voltage

Supply

Input

Logic

CPO

CP+

CP–

OPO

OP+

OP−

GND

OUHLogic

High side

Low side

VSH

GNDH

OUL

VSL

GNDL

InL

InH

−

UVLC

Delay

Com.

Level-

shifter

FIGURE 20.23 EUPEC EICE-Driver incorporating monolithic coreless transformer technology allowing for a substantial reduction in the device size.

Another manufacturer of isolated gate driver modules is

Eupec semiconductor [10]. These modules employ a mono-

lithically integrated coreless planar transformer design in a

gate driver IC. This allows it to be substantially smaller and

cheaper [6]. These modules are operational to 60 kHz with

output current capabilities in excess of 2 A peak. The EUPEC

EICE-Driver (2ED020I12-F) is shown in Fig. 20.23.

20.5.2 Non-isolated Electronic Level

Shifted Drivers

The International Rectiﬁer Corporation has developed a family

of low cost gate driver ICs for the power electronics industry

[11]. International Rectiﬁers, gate driver ICs (Fig. 20.24) utilize

electronic level shifting for conveying the switching signal to

556 I. Khan

28 Lead SOIC

28 Lead PDIP

44 Lead PLCC w/o 12 leads

14 Lead PDIP

IR2110/IR2113

14 Lead PDIP

w/o Lead 4

IR2110-1/IR2113-1

16 Lead PDIP

w/o leads 4 & 5

IR2110-2/IR2113-2

16 Lead SOIC

IR2110S/IR2113S

FIGURE 20.24 International Rectiﬁers IR2113 (Left) and the IR2233 (Right) high voltage gate driver ICs.

the high-side switch and a bootstrap technique for the gener-

ation of a ﬂoating supply. These ICs are relatively inexpensive

and are being used mainly in the low cost consumer elec-

tronics industry. Application include Electronic Ballasts and

low to medium power inverter applications such as variable

speed drives and DC to single-phase AC backup inverter sys-

tems. Available in a single-channel, half-bridge, or three-phase

inverter driver IC, these devices require a minimum amount

of external components. Intelligent features like under voltage

lockout, shut down, and current sensing inputs are included

in the driver IC.

Functional diagrams of the IR2113 half-bridge driver and

the IR2233 (Three-phase full-bridge) driver are shown in

Fig. 20.25. The driver section of the IR2233 comprises a similar

structure as the IR2113’s bootstrap and level shifting circuitry.

Even though the IR range of driver ICs do not provide iso-

lation between the control and driver electronics, this device

is a low cost and very compact solution for low to medium

power applications where electrical noise and interference can

be managed by PCB circuit layout techniques. The IR2113

half-bridge driver is capable of sourcing and sinking 2 A peak

current to a capacitive load without external buffer circuitry.

This feature coupled with low propagation delay times makes

this device capable of operating at frequencies in excess of

100 kHz.

The IR2113 and IR2233 are capable of driving ﬂoating

loads (high-side switches) at 600 and 1200 V DC bus volt-

ages, respectively, making them suitable for most direct off-line

single-phase and three-phase inverter applications.

Other driver ICs available include POWEREX’s M57959L

and INTERSIL’s HIP2500 bridge driver.

20.5.3 High-speed Gate Drivers

In the quest for high conversion efﬁciency and high power

densities, the trend in power electronics is the move towards

high converter switching frequencies at increasingly higher

power levels. The main reason for this is to reduce the

size of the energy transfer and storage components such as

transformers, capacitors, and inductors. This results in a sub-

stantial reduction in component size and to a certain extent

also cost. Achieving the objectives of high speed and high

power switching requires fast switching power semiconduc-

tors. These devices are readily available, but as the device

die-size increases with voltage and current requirements, so

does its gate drive power requirements. This is due to an

increase in device input capacitance as the die-size is increased.

Low impedance gate driver electronics capable of delivering

gate peak drive currents to large die-size IGBTs and power

MOSFETs in excess of 8 A are often required. In order to meet

these requirements, driver ICs must be able to deliver these

large pulse currents efﬁciently at high switching frequencies.

This is mainly due to a limitation in the maximum allowable

device power dissipation. Device package power dissipation is

typically limited to around 1.5 W for a DIP-14 IC without heat

sinking. Many manufacturers have high speed driver solutions

which realize drivers with low output impedances such as the

TC4427 and the TC4422. These devices are available as sin-

gle or dual channel drivers capable of delivering up to 9 A

peak to a 1–10 nF capacitive load with a typical rise time of

30–50 ns. A table of available high-speed drivers from TelCom

Semiconductor Inc. is shown in Table 20.3 [12].

20.5.4 Resonant Gate Drivers

High speed gate driving requires the driver and power semi-

conductor device impedance to be as low as possible. It is

also adversely affected by parasitic inductance and capacitance

caused by incorrect layout of the gate driver circuit. Some-

times these unavoidable parasitic elements tend to increase

the gate drive loop impedance, (between the driver and the

power semiconductor device) resulting in a limitation of the

peak current drawn by the device gate. This limits the maxi-

mum achievable switching speed of the device. It also results

in uncontrolled voltage excursions on the device gate, making

high-speed driver circuit design quite challenging.

Resonant gate drivers utilize these parasitic elements in

the gate drive loop by virtue of a controlled series resonant

20 Gate Drive Circuitry for Power Converters 557

International

Rectifier

Functional Block Diagram

INPUT

SIGNAL

GENERATOR

INPUT

SIGNAL

GENERATOR

INPUT

SIGNAL

GENERATOR

FAULT

LOGIC

CURRENT

COMPARATOR

UNDER

VOLTAGE

DETECTOR

CURRENT

AMP

RESET

SET

LATCH

DETECTOR

LATCH

DRIVER

DETECTOR

LATCH

DETECTOR

PULSE

GENERATOR

LEVEL

SHIFTER

PULSE

GENERATOR

LEVEL

SHIFTER

PULSE

GENERATOR

LEVEL

SHIFTER

COM

LO3

HO3

HO2

HO1

LO2

LO1

FAULT

FLT-CLR

0.5V

CAO

TRIP

CA–

CA+

LIN3

LIN2

LIN1

HIN1

HIN2

HIN3

–

IR2133/IR2135/IR2233/IR2235

IR2110/IR2113

Functional Block Diagram

International

Rectifier

DETECT

PULSE

FILTER

PULSE

GEN

LEVEL

SHIFT

LEVEL

SHIFT

LEVEL

SHIFT

DELAY

LIN

HIN

COM

FIGURE 20.25 Functional diagrams of the IR2113 half-bridge driver and the IR2233 three-phase full-bridge driver.

mode of operation. The resonant circuit comprises the driver

resistance (which is small), the device input capacitance, and

its internal gate electrode resistance. By the addition of a

deﬁned amount of extra inductance to the gate drive loop,

the voltage swing and switching speed of the gate driver can

be controlled. A typical resonant gate driver for large die-

power semiconductors has been developed by Turboswitchers

Inc. A resonant gate driver called the TD-000, employs a

patented low loss capacitance driver circuit topology, that

reduces the driver power losses to less than half that of

conventional high-speed drivers, at all switching frequencies.

Operational from a 5 V supply, the driver is capable of high-

switching speeds and generates a gate drive voltage swing

of up to +24 to −19 V, depending on the inductance val-

ues used. Performance tests show that the TD-000 driver is

capable of switching a 500 V/32 A power MOSFET supplied

from a 400 V DC bus, in less than 3 ns. This switching speed

allows for power electronic converters operating well into the

MHz range. Detailed device application information can be

found in [13].

558 I. Khan

TABLE 20.3 High current high-speed driver ICs available from TelCom semiconductor

Device no. Drive

current

(Peak)

Output number and type Rated

load

(pF)

Rise time

@ rated

load

(ns)

Fall time

@ rated

load

(ns)

Rising edge

propogation

delay

(ns)

Falling edge

propogation

delay

(ns)

Latch-up

proof

Input

protected

to 5V

below

Gnd rail

Inverting Non-inverting

TC1426 1.2 A Dual – 1000 30 20 55 80 Yes No

TC1427 1.2 A – Dual 1000 30 20 55 80 Yes No

TC1428 1.2 A Single Single 1000 30 20 55 80 Yes No

TC4426 1.5 A Dual – 1000 25 25 33 38 Yes Yes

TC4427 1.5 A – Dual 1000 25 25 33 38 Yes Yes

TC4428 1.5 A Single Single 1000 25 25 33 38 Yes Yes

TC4423 3.0 A Dual – 2200 25 25 33 38 Yes Yes

TC4424 3.0 A – Dual 2200 25 25 33 38 Yes Yes

TC4425 3.0 A Single Single 2200 25 25 33 38 Yes Yes

TC4420 6.0 A Single Non-invert 4700 40 35 50 55 Yes Yes

TC4429 6.0 A Single Inverting 4700 40 35 50 55 Yes Yes

TC4421 9.0 A Single – 10,000 50 48 30 33 Yes Yes

TC4422 9.0 A Single 10,000 50 48 30 33 Yes Yes

TC4469 1.2 A – Quad – 1000 30 30 35 35 Yes Yes

TC4468 1.2 A – Quad AND – 1000 30 30 35 35 Yes Yes

TC4467 1.2 A – Quad NAND – 1000 30 30 35 35 Yes Yes

20.6 Current and Future Trends

Integrated power electronic solutions have become the current

trend for applications with the development of Smart Power

modules. These modules contain the entire inverter semicon-

ductor stack as well as fully integrated gate drive circuitry.

This technology eliminates the challenges of inverter and gate

drive design and allows for fast turnaround times in new prod-

uct development through rapid prototyping. These devices are

available for low to medium power applications and have an

operating frequency limit of about 20 kHz.

Power semiconductor device manufacturers are constantly

developing better die-structures for their power MOSFETs

and IGBTs. This is to reduce device input capacitance, result-

ing in lower gate drive power requirements at high switching

speeds.

20.7 Summary

The aim of this chapter was to expose the reader to the basic

concepts, circuits, and technologies for gate drives in power

converters. The main focus has been on voltage-controlled

devices like power MOSFETs and IGBTs. Once the reader

gains a basic understanding of the concepts and available solu-

tions, detailed design information on gate drivers can be found

on the device manufacturer’s website. This is usually found

under technical application notes or technical white papers.

Other useful websites for power electronic design forums and

application-speciﬁc information can be found in [2, 14]. The

circuits presented can be adapted for the driving of other

devices such as SCRs and power BJTs.

References

1. N. Mohan, T. Undeland, and W. Robbins, “Power Electronics:

Converters, Applications and Design”, Wiley, Brisbane, 1989.

2. www.powerdesigners.com

3. D. R. H. Carter, “Aspects of High Frequency Half-Bridge Circuits”,

PhD Thesis, Cambridge University, September 1996.

4. S. Clement and A. Dubhashi, “HV Floating MOS-Gate Driver IC”,

Integrated circuit designers manual.

5. Application Note, “Hints and Applications” Design manual,

Chapter 3, Semikron Corporation.

6. M. Munzer, W. Ademmer, B. Strazalkowski, and K. T. Kaschani,

“Coreless Transformer a New Technology for Half Bridge Driver

IC’s”, application note, 2005, www.eupec.com.

7. I. de Vries, “High Power and High Frequency Class-DE Inverters”,

PhD Thesis, Department of Electrical Engineering, University of Cape

Town, August 1999.

8. Application Note AN-937, “Gate Drive Characteristics and Require-

ments for HEXFET Power MOSFETs”, www.irf.com.

9. Data sheet, SKHI22, www.semikron.com.

10. www.eupec.com

11. www.irf.com

12. Application Note 30, “Matching MOSFET drivers to MOSFETs”,

TelCom Semiconductor Inc.

13. I. de Vries, “Using Turbodriver-000”, application note, February

2002, www.turboswitchers.com.

14. www.smpstech.com

Power Electronics in Capacitor

Charging Applications

William C. Dillard, Ph.D.

Archangel Systems, Incorporated

1635 Pumphrey Avenue Auburn,

Alabama, USA

21.1 Introduction .......................................................................................... 559

21.2 High Voltage DC Power Supply with Charging Resistor................................. 560

21.3 Resonance Charging ................................................................................ 561

21.4 Switching Converters............................................................................... 562

References ............................................................................................. 564

21.1 Introduction

Conventional dc power supplies operate at a given dc out-

put voltage into a constant or near constant load. Pulse loads

such as lasers, ﬂashlamps, railguns, and radar, however, require

short but intense bursts of energy. Typically, this energy is

stored in a capacitor and then released into the load. The

rate at which the capacitor is charged and discharged is called

the repetition rate, T, and may vary from 0.01 Hz for large

capacitor banks to a few kHz for certain lasers. Recharging the

capacitor voltage to a speciﬁed voltage is tasked to a capacitor

charging power supply (CCPS). The role of power electron-

ics devices, topologies, and charging strategies for capacitor

charging applications is presented in this chapter.

Figure 21.1 shows the voltage across the energy storage

capacitor connected to the output of a CCPS. As seen in this

ﬁgure, the CCPS has three modes of operation. The ﬁrst mode

is the charging mode in which the capacitor is charged from

an initial voltage of zero to a speciﬁed ﬁnal voltage. The dura-

tion of the charging mode is determined by the capacitance of

the energy storage capacitor and the rate at which the CCPS

delivers energy. The next mode of operation is the refresh

mode, which can be considered a “standby mode” where the

stored energy is simply maintained. When the output volt-

age drops below a predetermined value, the CCPS should

turn on and deliver the energy necessary to compensate for

capacitor leakage. Since refreshing is lost energy, the dura-

tion of the refresh mode should be as brief as possible. Issues

that lead to non-zero refresh times include safety margins for

worst-case charging and discharging mode times and SOA

requirements of switching devices. The ﬁnal mode of operation

is the discharge mode where the load is actively discharging the

capacitor. The CCPS does not supply any energy to the load

in this mode. The amount of time the CCPS remains in this

mode is determined by how quickly the load can discharge the

capacitor.

The instantaneous output power for a CCPS varies over a

wide range in comparison to a conventional dc power sup-

ply which supplies a near constant power to its load. This is

shown in Fig. 21.2; the output power for the pulsed power

load is drawn as linear for illustration purposes only. The

charging mode is characterized by high peak power. At the

beginning of this mode, the output power is zero (i.e. there

is no voltage present but current is ﬂowing). Thus, the load

capacitor is equivalent to a short circuit. Additionally, at the

end of the charging mode, the output power is again zero (i.e.

there is an output voltage present but no current is ﬂowing).

Now the load capacitor is equivalent to an open circuit. The

refresh mode is typically a low power mode, because the cur-

rent required to compensate for capacitor leakage is small. The

CCPS does not supply any power during the discharge mode

when the energy storage capacitor is being discharged by the

pulsed load.

The average output power for a CCPS depends on the dis-

charge mode energy and the repetition rate of the load. It is

maximum when the energy storage capacitor is discharged

at the end of the charging mode (large voltage and current),

which corresponds to operation without a refresh mode. Since

the CCPS power is far from constant, the rating of a CCPS

is often given in kJ/s instead of kW. The kJ/s rating can be

559

560 W. C. Dillard

Output Voltage

time

Charging

Mode

Refresh

Mode

Discharge

Mode

Repetition Rate = 1/T

FIGURE 21.1 Three modes of operation of a capacitor charging power supply.

Power

time

Pulsed Power Load Constant Power Load

Charging

Mode

Refresh

Mode

Discharge

Mode

FIGURE 21.2 Power requirements for pulsed power and constant power loads.

written as,

kJ/s =

LOAD

where W

LOAD

is the energy delivered to the load per charging

cycle and T is the repetition rate. In the optimum case with no

refresh and instantaneous discharge, the kJ/s rating is limited

to how fast a particular capacitor can be charged by its speciﬁed

voltage.

21.2 High Voltage DC Power Supply

with Charging Resistor

In this technique, the energy storage capacitor is charged by

a high voltage dc power supply through a charging resistor

as illustrated in Fig. 21.3. The charging mode ends when the

capacitor voltage equals the output voltage of the power sup-

ply. The capacitor is continually refreshed by the power supply.

During the discharge mode, the charging resistor isolates the

High Voltage

DC Power

Supply

Pulse Load

FIGURE 21.3 High voltage dc power supply and charging resistor.

power supply from the pulse load. The advantages of this

technique are its simplicity, reliability, and low cost.

The major disadvantage of this technique is its poor efﬁ-

ciency. In the charging mode, the energy dissipated in the

charging resistor is equal to the energy stored in the capacitor

in the ideal case; therefore the maximum efﬁciency is 50%. As

a result, this technique is utilized only in applications where

the charge rate is low, i.e. 200 J/s. Another disadvantage of this

technique is related to the charging time, which is determined

by the RC time constant. Some laser applications require that

the output voltage be within 0.1% of a target voltage. For this

technique, more than ﬁve time constants are required for the

capacitor voltage to meet this voltage speciﬁcation.