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21 Power Electronics in Capacitor Charging Applications 561

21.3 Resonance Charging

The basic resonance charging technique is illustrated in

Fig. 21.4. An ac input voltage is stepped up with a transformer,

rectiﬁed, and ﬁltered with capacitor C

to produce a high dc

voltage V

. In this circuit, C

is much greater than C

. Thyris-

tor T

is gated and current ﬂows through the inductor and

diode D

is transferring energy from C

to C

. The voltage

v(t) and current i(t) are described by the following equations

assuming that C

 C

. The charge time, t

, for this circuit

can be calculated by ﬁnding the time at which the current,

described by Eq. (21.2), reaches zero and is given below. The

voltage v(t) has a value of 2V

at the end of the charging mode.

v(t ) = V

(1 −cos ω

t) (21.1)

i(t) = V



sin ω

t (21.2)

√

(21.3)

= π



(21.4)

Even though this technique is simple and efﬁcient, it is not

without its limitations. A high voltage capacitor with a large

capacitance value is needed for C

, which increases the cost.

A single thyristor is shown in Fig. 21.4. Multiple thyristors

connected in series or a thyratron may be required depending

on the voltage level. The repetition rate of the pulse load should

be such that C

is fully charged and i(t) has reached zero before

Rectifier Pulse Load

1:N

–

v(t)

–

i(t)

FIGURE 21.4 Resonance charging.

Rectifier Pulse Load

1:N

–

v(t)

–

Control

FIGURE 21.5 Resonance charging with de-queing.

the load discharges to prevent latch up of T

. It is not possible

for this circuit to operate in the refresh mode because of the

switch characteristics. Therefore, v(t) will drift due to capacitor

leakage. The charge time is a function of the circuit parameters

and will drift as they change with temperature or due to aging.

Since all of the energy stored in C

is transmitted from C

in a single pulse, it can be difﬁcult to achieve precise voltage

regulation with the resonance charging technique. However,

regulation can be improved with the addition of a de-queing

circuit as shown in Fig. 21.5. The voltage v(t) is monitored with

a sensing network. Just before v(t) reaches the desired level,

thyristor T

is ﬁred which terminates the charging mode. The

remaining energy stored in the inductor is dissipated in R. The

addition of the de-queing circuit reduces the circuit efﬁciency

and increases the circuit complexity and cost, but still does not

enable a refresh mode to compensate for capacitor leakage.

Boost charging, a variation on the resonance charging tech-

nique, is shown in Fig. 21.6 [1]. An extra switch is added to

the circuit of Fig. 21.4 allowing energy to be stored in both C

and L. This can be modeled as an increase in the voltage gain

of the CCPS. With switches S

and S

closed, the current i(t)

is given by

i(t) =

t (21.5)

At some time t = t

is opened, and the current is now

described by

i(t) = I

cos ω

t +V



sin ω

t (21.6)

562 W. C. Dillard

Rectifier Pulse Load

1:N

–

v(t)

–

i(t)

FIGURE 21.6 Boost charging.

where

(21.7)

is the inductor current initial value at t

. The voltage v(t)is

then

v(t ) = I



sin ω

t +V

(

1 −cos ω

)

(21.8)

The time required for the current to reach zero and for the

voltage v(t) to reach its peak value can be calculated from

tan

(

)

=−



=−ω

(21.9)

This is also the charge time, t

, or the length of the charging

mode. Note from equation (21.9) that the charge time depends

on t

, which is the on-time of switch S

. In addition, the

peak capacitor voltage is a function of t

. The peak capacitor

voltage is limited to 2V

without S

; voltage gains as high as

20 are possible with the addition of S

[1].

The switching elements in Figs. 21.4 and 21.5 are realized

with thyristors. Simple switches are shown for the boost-

charging technique in Fig. 21.6. Switch S

could be imple-

mented with a thyristor. The boost capability provided by

switch S

is best realized with a gate-controlled semiconductor

device such as a GTO or an IGBT.

21.4 Switching Converters

The poor efﬁciency when charging a capacitor through a resis-

tor from a high-voltage power supply limits its application to

low charging rates. In the resonance-charging concepts, the

energy is transferred to the load capacitor in a single pulse,

and it is not possible to compensate for capacitor leakage.

Energy storage capacitors may be charged utilizing the same

power electronic technology that has been applied in switch-

ing converters for constant power loads. Instead of charging

the energy storage capacitor with a single pulse, switching con-

verters can charge the capacitor with a series of pulses or pulse

train. The peak current is reduced when charging with a series

of pulses, thus improving the efﬁciency of the charging pro-

cess. In addition, soft-switching techniques may be employed

in the switching converter to increase the efﬁciency. The reg-

ulation of the output voltage is also improved with the pulse

train, because the energy is passed to the energy storage capac-

itor as small packets. Common control techniques such as

pulse-width modulation can be used to control the size of the

energy packet. This capability to control the size of the energy

packet permits the CCPS to operate in the refresh mode and

compensate for capacitor leakage. As a result, the CCPS may

operate over a broad range of load repetition rates and still

maintain tight output voltage regulation during refresh. Dur-

ing the refresh mode, energy lost due to capacitor leakage may

be replaced in a burst fashion [2] or in a continuous fashion

similar to trickle charging a battery [3].

In the switching converter, semiconductor switches may be

operated on the low side of the transformer permitting the use

of MOSFETs or IGBTs in the CCPS. Since the CCPS begins

the charging mode with a short circuit across its output, the

switching converter must be capable of operating under this

severe load condition. This may require the implementation

of a current limiting scheme in the converter control circuit.

One switching converter, topology candidate is the series

resonant converter in Fig. 21.7. Note that the MOSFETs and

resonant components L

and C

are connected on the low

voltage side of the transformer. Only the rectiﬁer diodes and

energy storage capacitor must have high voltage ratings. When

the output rectiﬁer is conducting, the energy storage capacitor

is connected in series with the resonant capacitor C

. For

a transformer turns ratio of 1:N, reﬂecting C

through the

transfomer yields a capacitance of N

. Since N is typically

large, this reﬂected capacitance is much larger than C

, thus

the resonant frequency, which is deﬁned in Eq. (21.10), is not

affected by C

. For high voltage, high-frequency operation, the

leakage inductance of the transformer may be utilized as L

Thus, the resonant frequency can be expressed as,

√

((NC

)/(C

+NC

))

(21.10)

One characteristic of this converter, which makes it attrac-

tive for capacitor charging, is the ability to operate under the

short circuit conditions present at the beginning of the charg-

ing mode. The voltage across C

is zero at the beginning of

21 Power Electronics in Capacitor Charging Applications 563

T2T3

1:N

FIGURE 21.7 Series resonant converter.

this mode. The current in the switches is limited by the input

voltage and impedance Z

as is deﬁned in Eq. (21.11).



(21.11)

Another method for current limiting is to vary the ratio of

, the switching frequency of the MOSFETs, and the resonant

frequency, f

, which is ω

/2π. This effectively controls the ﬂow

of energy from the source to C

. The ratio f

may be set to a

low value at the beginning of the charging mode and increased

toward one as the voltage across C

builds up. This limits the

current when the voltage across C

is low and allows increased

energy transfer as the voltage approaches the target voltage.

The disadvantage of this approach is that variable frequency

operation complicates device and component selection and

degrades EMI/EMC performance of the CCPS.

The ﬂyback converter, shown in Fig. 21.8, also may be

utilized for capacitor charging applications [4, 5]. When the

MOSFET is turned on, current builds up in the primary wind-

ing storing energy in the magnetic ﬁeld. When it reaches a

speciﬁed level, the MOSFET is turned off and energy is trans-

ferred from the magnetic ﬁeld to C

. This energy transfer is

terminated when the MOSFET is turned on again.

In cases where precise output regulation is not required and

the packet energy is low, the diode in Fig. 21.8 can be replaced

with a Zener diode with a Zener voltage of

= nV

O,nom

where V

O,nom

is the nominal value of V

[6]. Once V

O,nom

is reached, the next energy packet will force the diode into

a brief period of breakdown with the excess energy partially

N1:N2

FIGURE 21.8 Flyback converter.

recycled to the input circuit. If the packet energy is small

enough, the breakdown is not destructive and V

is limited to

= V

O,nom

+V

O,

where V

O,

is the small excess voltage caused by the last

energy packet.

Sokal and Redl [7] have investigated different control

schemes for charging capacitors using the ﬂyback converter.

Their recommendation is to charge C

with current pulses

which are nearly ﬂat-topped. This strategy results in higher

average current for a given peak current. The capacitor is

charged faster because the charge delivered to it during a pulse

is directly proportional to the average current. This desired

pulse shape is achieved by turning on the MOSFET to termi-

nate the transfer of energy to C

soon after the MOSFET is

turned off, which increases the switching frequency. When the

564 W. C. Dillard

1:N

FIGURE 21.9 Ward converter.

primary current rises to a preset minimum level, the MOSFET

is again turned off. This switching strategy is essentially hys-

teretic current mode control, where the switch current is

limited between two preset bounds.

Another converter for capacitor charging applications is the

Ward converter [8–10] shown in Fig. 21.9. When the MOSFET

is on, energy is stored in the inductor and capacitor C

trans-

fers energy to the energy storage capacitor C

and capacitor C

The energy stored in the inductor is transferred to C

when

the MOSFET is off. The leakage inductance of the transformer

and C

resonate producing a sinusoidal current which ﬂows

in the primary winding of the transformer and the MOSFET.

When the primary current reaches zero and starts negative,

the diode turns on which allows the MOSFET to be turned off

efﬁciently at zero current.

In some converter operating conditions, the voltage across

is very small because most of the energy has been trans-

ferred from C

to C

. The energy stored in C

may be too

small to ensure zero-current turn off of the MOSFET. In this

case, the energy stored in C

helps to ensure that the ampli-

tude of the current is large enough for zero-current turn off of

this device.

References

1. P. K. Bhadani, “Capacitor-Charging Power Supply for Laser Pulsers

Using a Boost Circuit,” Review of Scientiﬁc Instruments, Vol. 60,

No. 4, April 1989, pp. 605–607.

2. A. C. Lippincott and R. M. Nelms, “A Capacitor-Charging

Power Supply Using a Series-Resonant Topology, Constant

On-Time/Variable Frequency Control, and Zero Current Switching,”

IEEE Transactions on Industrial Electronics, Vol. 38, No. 6,

December 1991, pp. 438–447.

3. B. E. Strickland, M. Garbi, F. Cathell, S. Eckhouse, and M. Nelms,

“2 kJ/sec 25-kV High-Frequency Capacitor-Charging Power Sup-

ply Using MOSFET Switches,” Proceedings of the 1990 Nineteenth

Power Modulator Symposium, June 1990.

4. R. L. Newsom, W. C. Dillard, and R. M. Nelms, “Digital Power-

Factor Correction for a Capacitor-Charging Power Supply,” IEEE

Transaction on Industrial Electronics, Vol. 49, No. 5, October 2002,

pp. 1146–1153.

5. F. P. Dawson and S. B. Dewan, “A Subresonant Flyback Converter

for Capacitor Charging,” Proceedings of the Second Annual IEEE

Applied Power Electronics Conference, March 1987, pp. 274–283.

6. W. C. Dillard and R. M Nelms, “A Digitally-Controlled, Low-

Cost Driver for Piezoceramic Flight Control Surfaces in Small

Unmanned Aircraft and Munitions,” Seventeenth Annual Applied

Power Electronics Conference and Exposition, 2002, APEC ’02,

pp. 1154–1160.

7. N. O. Sokal and R. Redl, “Control Algorithms and Circuit Designs for

Optimally Flyback-Charging an Energy-Storage Capacitor (e.g. for

ﬂash lamp or deﬁbrillator),” IEEE Transactions on Power Electronics,

Vol. 12, No. 5, September 1997, pp. 885–894.

8. M. A. V. Ward, “DC to DC Converter Current Pump,” U.S. Patent

Number 4,868,730, September 1989.

9. G. C. Chryssis, “High-Frequency Switching Power Supplies: Theory

and Design,” McGraw-Hill Publishing, New York, 1989.

10. R. M. Nelms and J. E. Schatz, “A Capacitor Charging Power Sup-

ply Utilizing a Ward Converter,” IEEE Transactions on Industrial

Electronics, Vol. 39, No. 5, October 1992, pp. 421–428.

Electronic Ballasts

J. Marcos Alonso, Ph.D.

University of Oviedo, DIEECS -

Tecnologia Electronica, Campus

de Viesques s/n, Ediﬁcio de

Electronica, 33204 Gijon,

Asturias, Spain

22.1 Introduction .......................................................................................... 565

22.1.1 Basic Notions • 22.1.2 Discharge Lamps • 22.1.3 Electromagnetic Ballasts

22.2 High Frequency Supply of Discharge Lamps ................................................ 571

22.2.1 General Block Diagram of Electronic Ballasts • 22.2.2 Classiﬁcation of Electronic

Ballast Topologies

22.3 Discharge Lamp Modeling........................................................................ 575

22.4 Resonant Inverters for Electronic Ballasts .................................................... 578

22.4.1 Current-fed Resonant Inverters • 22.4.2 Voltage-fed Resonant Inverters

• 22.4.3 Design Issues

22.5 High Power Factor Electronic Ballasts......................................................... 586

22.5.1 Harmonic Limiting Standards • 22.5.2 Passive Solutions • 22.5.3 Active Solutions

22.6 Applications........................................................................................... 589

22.6.1 Portable Lighting • 22.6.2 Emergency Lighting • 22.6.3 Automotive Lighting

• 22.6.4 Home and Industrial Lighting • 22.6.5 Microprocessor-based Lighting

Further Reading ..................................................................................... 590

22.1 Introduction

Electronic ballasts, also called solid-state ballasts, are those

power electronic converters used to supply discharge lamps.

The modern age of electronic ballasts began with the intro-

duction of power bipolar transistors with low storage time,

allowing to supply ﬂuorescent lamps at frequencies of several

kilohertz and increasing lamp luminous efﬁcacy by operating

at these high frequencies. Later, electronic ballasts became very

popular with the development of low cost power MOSFETs,

whose unique features make them very attractive to implement

solid-state ballasts. The main beneﬁts of electronic ballasts are

the increase in the lamp and ballast overall efﬁciency, increase

in lamp life, reduction of ballast size and weight, and improve-

ment in lighting quality. This chapter attempts to give a general

overview about the more important topics related to this type

of power converters.

22.1.1 Basic Notions

Discharge lamps generate electromagnetic radiation by means

of an electric current passing through a gas or metal vapor.

This radiation is discrete, as opposed to the continuous

radiation emitted by an incandescent ﬁlament. Figure 22.1

shows the electromagnetic spectrum of an electric discharge,

which consists of a number of separate spectral lines.

As can be seen in Fig. 22.1, only the electromagnetic radi-

ation emitted within the visible region (380–780 nm) of the

radiant energy spectrum is useful to provide lighting. The total

power in watts emitted by an electric discharge can be obtained

by integrating the spectral energy distribution. However, this

is not a suitable parameter to measure the amount of light

emitted by a discharge lamp.

The human eye presents different responses to the differ-

ent types of electromagnetic waves within the visible range. As

illustrated in Fig. 22.2, there exist two response curves. First,

the photopic curve, also called V (λ), is the characteristic used

to represent the human eye behavior under normal illuminat-

ing level conditions or daylight vision. Second, the scotopic

curve V



(λ) is the response of the human eye for situations

with low illuminating levels, also known as nocturnal vision.

The reason for this different behavior is physiologic. The

human eye consists of two types of photoreceptors: rods and

cones. Rods are much more sensitive at low lighting levels than

cones, but they are not sensitive to the different light colors.

On the other hand, cones are responsible for normal color

565

566 J. M. Alonso

400 500 600 700

λ (nm)

P (λ)

VISIBLE LIGHT: 380–780 nm

−12

−6

−4

−2

110

−8

−10

UV IR

λ (nm)

Microwave TV & RadioX-Rays

−14

Gamma

Rays

Cosmic

Rays

FIGURE 22.1 Spectral energy distribution of an arc discharge and radiant energy spectrum.

0.2

0.4

0.6

0.8

380 480 580 680 780

Photopic

Scotopic

λ (nm)

FIGURE 22.2 Spectral luminous efﬁciency functions for standard pho-

topic and scotopic observers.

vision at higher lighting levels. Normally, only the photopic

function is considered in lighting design and used to calibrate

photometers.

Since, the human eye responds in different ways to the dif-

ferent wavelengths or colors, the output power of a lamp,

measured in watts, is no longer applicable to represent the

amount of light generated. Thus, a new unit is used to incor-

porate the human eye response, which is called lumen. The

total light output of a lamp is then measured in lumens and

it is known as lamp luminous ﬂux. The lamp luminous ﬂux is

obtained by integrating the radiant power as follows:

 = K



780nm

380nm

P(λ)V (λ)dλ (22.1)

where K

represents the maximal spectral luminous efﬁ-

cacy, which is equal to 683 lumens/watt (at λ = 555 nm)

for photopic vision and 1700 lumens/watt (at λ = 507 nm)

for scotopic vision. The standard photopic and scotopic

functions were deﬁned by the International Commission on

Illumination (CIE) in 1924 and 1951, respectively.

The measurement of the lamp total luminous ﬂux results is

very useful to know whether or not the lamp is working prop-

erly. At the laboratory, the measurement of the lamp total

luminous ﬂux is performed by means of an integrating sphere

and using the substitution method. The integrating sphere,

also known as Ulbricht photometer, is internally coated with

a perfectly diffusing material. Thus, the sphere performs the

integral in Eq. (22.1) and the illuminance on the internal sur-

face is proportional to the total luminous ﬂux. A photometer

with a V (λ) ﬁlter is placed so that the internal illuminance can

be measured, and a bafﬂe is placed to avoid direct illumina-

tion of the photometer probe by the lamp. The measurement

is made in two steps, one with the lamp under test in place and

the other with a standard lamp of known total luminous ﬂux.

From the two measurements, the total luminous ﬂux of the

lamp under test is deduced by linear relationship. Figure 22.3

illustrates an integrating sphere photometer.

22 Electronic Ballasts 567

Lamp

Baffle

Photometer with

V(λ) probe

Perfectly Diffusing

Coating

FIGURE 22.3 Integrating sphere used for measurement of the lamp

total luminous ﬂux.

An important parameter related to the supply of discharge

lamps is the luminous efﬁcacy. Luminous efﬁcacy is deﬁned as

the rate of the lamp total luminous ﬂux to the total electric

power consumed by the lamp, usually expressed in lumens

per watt. The luminous efﬁcacy of a discharge lamp can be

increased by proper designing of the electronic ballast, which

ﬁnally results in energy saving.

22.1.2 Discharge Lamps

Basically, discharge lamps consists of a discharge tube inside

which the electric energy is transformed into electromagnetic

radiation. The discharge tube is made of a transparent or

translucent material with two sealed-in electrodes placed at

both ends, as shown in Fig. 22.4. The discharge tube is ﬁlled

with an inert gas and a metal vapor. The electrodes gener-

ate free electrons, which are accelerated by the electrical ﬁeld

existing in the discharge. These accelerated electrons collide

with the gas atoms, having both elastic and inelastic collisions

depending on the electron kinetic energy. The basic processes

inside the discharge tube are illustrated in Fig. 22.4. They are

as follows:

1. Heat generation. When the kinetic energy of the electron

is low, an elastic collision takes place and only a small part of

Heat generation Energy radiation Ionization

Electrode

Electrons

FIGURE 22.4 Basic processes inside the discharge tube.

the electron energy is transferred to the gas atom. The result

of this type of collisions is an increase in the gas tempera-

ture. In this case, the electrical energy is consumed to produce

heat dissipation. However, this is also an important process

because the discharge has to set in its optimum operating

temperature.

2. Gas atom excitation. Some electrons can have a high

kinetic energy so that the energy transferred in the collision

is used to send an electron of the gas atom to a higher orbit.

This situation is unstable and the electron tends to recover its

original level, then emitting the absorbed energy in the form

of electromagnetic radiation. This radiation is used to directly

generate visible light or, in other case, ultraviolet radiation is

ﬁrst generated and then transformed into visible radiation by

means of a phosphor coating existing in the inside wall of the

discharge tube.

3. Gas atom ionization. In some cases, electrons have gained

such high kinetic energy that during the collision with the gas

atom, an electron belonging to the gas atom is freed, resulting

in a positive charged ion and a free electron. This freed electron

can play the same roles as those generated by the electrodes.

This process is specially important during both discharge igni-

tion and normal running, because ionized atoms and electrons

are necessary to maintain the electrical current through the

lamp.

The number of free electrons in the discharge can increase

rapidly due to continuing ionization, producing an unlim-

ited current and ﬁnally a short-circuit. This is illustrated in

Fig. 22.5, which shows how the voltage–current characteristics

of a gas discharge exhibits a negative differential resistance.

Therefore, in order to limit the discharge current, the use of

an auxiliary supply circuit is mandatory. This circuit is called

ballast.

Focusing on discharge lamps, the complete stabilization

process consists of two main phases:

1. Breakdown phase. Most of the gases are very good insu-

lators and an electric discharge is only possible if a sufﬁcient

568 J. M. Alonso

− I

− V

< 0

FIGURE 22.5 Voltage–current characteristics of a gas discharge.

concentration of charged particles is present. Normally, a high

voltage is used to provide electricity carriers and to initi-

ate the discharge. The minimum voltage applied to initiate

the discharge is called starting voltage. The starting voltage

mainly depends on the type of gas, gas pressure, and distance

between electrodes. Figure 22.6a represents the starting volt-

age as a function of the product gas pressure by electrode

distance, for different gases. These functions are known as

Paschen curves.

Usually, auxiliary inert gases are used to decrease the starting

voltage. Particularly, there exist some special inert gas mix-

tures, presenting a very low starting voltage, which are called

Penning mixtures. These penning mixtures are very often used

1 10 100

100

200

500

1000

2000

(V)

Ne + 0.1% Ar

p·l

(Pa·m)

0 1.0 2.0 3.0 4.0

100

150

Time (minutes)

I - Current

V- Voltage

P- Power

Φ- Total Luminous Flux

(a) (b)

FIGURE 22.6 (a) Paschen curves for different inert gases and (b) stabilization curve of a discharge lamp.

as initial starting gas. Figure 22.6a shows a typical penning

mixture constituted by neon with 0.1% of argon.

2. Warm-up phase. Once the lamp is ignited, the collisions

between free electrons and atoms generate heat and discharge

temperature increases until reaching the normal operation

conditions. During this phase, the heat is used to evaporate

the metal atoms existing in the discharge tube and the emit-

ted electromagnetic radiation assumes the character of a metal

vapor discharge instead of an inert gas discharge. From the

electrical point of view, the discharge warm-up phase shows

initially low discharge voltage and high discharge current. As

long as more and more metal atoms are evaporated, the dis-

charge voltage increases and the discharge current decreases.

Finally, an equilibrium is reached at steady-state operation

with the normal values of voltage and current. The time con-

stant of the warm-up phase strongly depends on the lamp

type. It varies from seconds for ﬂuorescent lamps to minutes

for high-intensity discharge lamps.

Figure 22.6b illustrates some discharge waveforms during

the warm-up phase.

The basic elements used in lamps to generate radiation in

the visible part of the spectrum are sodium and mercury. The

former generates radiation directly within the visible part of

the spectrum; the latter generates radiation mainly in the ultra-

violet region, but this radiation can be easily transformed in

visible radiation by means of a phosphor coating on the inter-

nal wall of the discharge tube. Besides the element used, a very

important parameter related to the efﬁciency and richness of

the emitted radiation is the discharge pressure. For the sodium

and mercury elements, there exist two pressure values around

22 Electronic Ballasts 569

which the luminous efﬁcacy of the discharge is higher. The ﬁrst

is obtained at quite low-pressures, about 1 Pa, and the second

at higher pressures, around 10

Pa (1 at). This is the reason

why there exist two main types of discharge lamps:

1. Low-pressure discharge lamps. This type of lamps oper-

ates with pressures around 1 Pa and they feature low current

density inside the discharge and low power per unit of dis-

charge length. Therefore, these lamps present normally a quite

large discharge volume with low power rating. Most represen-

tative examples are low-pressure mercury lamps, also known

as ﬂuorescent lamps and low-pressure sodium lamps.

2. High-pressure discharge lamps. The operating pressure

in this type of lamps is around 10

Pa and higher, in order

to achieve a considerable increase in the luminous efﬁcacy

of the discharge. These lamps present a high-current density

in the discharge and a high power per discharge length ratio,

thus showing much smaller discharge tubes. Examples of these

lamps are high-pressure sodium lamps, high-pressure mercury

lamps, and metal halide lamps.

Finally, to characterize the light produced by a discharge

lamp, it is necessary to know two important concepts: the

correlated color temperature (CCT) and the color rendering

index (CRI).

The CCT is deﬁned as the temperature of the blackbody

radiator whose perceived color most closely resembles that of

the discharge lamp. The color of an incandescent body changes

as its temperature rises from deep red to orange, yellow, and

ﬁnally white. Thus, a cool white ﬂuorescent lamp has a CCT

around 3500 K and it is perceived as a white source of light,

whereas a high pressure-sodium lamp presents a CCT of about

2000 K and it appears as yellow.

The CRI of a light source is the effect that the source has on

the color appearance of the objects under it, when compared

to their appearance under a reference source of equal CCT.

The measurement gives a value lower than 100, and the higher

the CRI, the better the color rendition. For example, daylight

and incandescent lamps has a CRI equal to 100.

To conclude this introduction on the discharge lamps, some

comments regarding the most important types of discharge

TABLE 22.1 Comparison of different discharge lamps

Lamp Wattage (W) Luminous efﬁcacy (lm·W

−1

) Life (h) CCT (K) CRI

Fluorescent 4–100 62 20,000 4200 62

Compact ﬂuorescent 7–30 60–80 10,000 2700–5000 82

Low-pressure sodium 50–150 110–180 15,000 1800 <0

Mercury vapor 50–1000 40–70 24,000 4000–6000 15–50

Metal-halide 40–15,000 80–125 10,000 4000 65

HPS 35–1000 65–140 24,000 2000 22

HPS(Amalgam) 35–1000 45–85 10,000 2200 65

lamps will be given. Table 22.1 provides some additional data

on the different discharge lamps for comparison.

1. Fluorescent lamps. These lamps belong to the category of

low-pressure mercury vapor discharge lamps. The discharge

generates two main lines at 185 and 253.7 nm and other weak

lines in the visible range of the spectrum. A ﬂuorescent powder

on the inside wall of the discharge tube converts the ultraviolet

radiation into visible radiation, resulting in a broadband spec-

tral distribution and good color rendition. In these lamps, the

optimum mercury vapor pressure (which gives the maximum

luminous efﬁcacy) is 0.8 Pa. For the tube diameters normally

used, this pressure is reached at a wall temperature of about

◦

C, not much higher than typical ambient temperature. The

heat generated inside the discharge is sufﬁcient to attain the

required operating temperature without using an outer bulb.

However, this structure causes a great variation of the lamp

lumen output with the temperature, which is one important

drawback of the ﬂuorescent lamps. One solution to this prob-

lem is the addition of amalgams to stabilize the light output.

This is specially used in compact ﬂuorescent lamps.

2. Low-pressure sodium lamps. These lamps are the most

efﬁcient source of light. The reason is the almost monochro-

matic radiation that they generate, with two main lines at 589

and 589.6 nm, very close to the maximum human eye sensi-

tivity. Therefore, color rendition of these lamps is very poor,

however contrasts are seen more clearly under this light. This

is the reason for using these lamps in situations where the

recognition of objects and contours is essential for safety, such

as motorway bridges, tunnels, intersections, and so on. The

optimum pressure for the low-pressure sodium discharge is

about 0.4 Pa, attained in normal discharge tubes at a temper-

ature of 260

◦

C. An outer bulb is normally used to reach and

maintain this temperature.

3. High-pressure mercury vapor lamps. The increase in the

pressure of the mercury vapor produces a radiation richer

in spectral lines, some of them are in the visible part of

the spectrum (405, 436, 546, and 577/579 nm). This leads

570 J. M. Alonso

to an increase in the luminous efﬁcacy, reaching values of

40–60 lm·W

−1

at pressures 10

–10

Pa (1–100 at). These

lamps operate with unsaturated mercury vapor, which means

that all the mercury in the discharge tube has evaporated

and the number of mercury atoms per unit volume remains

constant. Thus, the operation of this type of lamps is more

independent of the temperature than most other discharge

lamps. One drawback of these lamps is the lack of spectral lines

in the long wavelengths (reds) of the spectrum, thus showing

low CRI. An increase of the color rendition can be obtained

by adding metal-halide compounds into the discharge tube, in

order to generate radiation all over the visible spectrum. These

lamps are known as metal-halide lamps.

4. High-pressure sodium lamps. This is a very popular

source of light due to its high luminous efﬁcacy and long

life. The increase in the sodium vapor pressure produces a

very widening spectrum, with good color rendition compared

to the low-pressure sodium lamps. This also leads to a lower

luminous efﬁcacy but it is still higher than the other high

intensity discharge lamps. Some of these lamps also incorpo-

rate mercury in the form of sodium amalgam to increase the

ﬁeld strength of the discharge, thus decreasing the discharge

current. A lower lamp current and a higher lamp voltage

allows to reduce the size and cost of the ballast. However,

the addition of sodium amalgam strongly reduces the life of

the lamp.

22.1.3 Electromagnetic Ballasts

Electromagnetic ballasts are commonly used to stabilize the

lamp at the required operating point by limiting the discharge

current. The operating point of the lamp is given by the inter-

section of both lamp and ballast characteristics, as shown in

Fig. 22.7. The ballast line is the characteristic which shows

the variation of the lamp power vs lamp voltage for a con-

stant line voltage, and can be measured during the warm-up

phase of the lamp. The lamp line is the characteristic which

gives the variation of the lamp power as a function of the

lamp voltage for different line voltages, and can be measured

by varying the line voltage. Some lamps, like high-pressure

sodium, exhibit a great variation of lamp voltage with changes

in the lamp wattage. Because of this behavior, trapezoids have

been established which deﬁne maximum and minimum per-

missible lamp wattage vs lamp voltage for purposes of ballast

design, as shown in Fig. 22.7.

Figure 22.8 shows basic electromagnetic ballast used to

supply low- and high-pressure lamps at line frequencies

(50–60 Hz). Figure 22.8a illustrates the typical circuit used

to supply ﬂuorescent lamps with preheating electrodes, which

basically uses a series inductor to limit the current through

the discharge. Initially the glow switch is closed and the short-

circuit current ﬂows through the circuit, heating the electrodes.

LAMP

WATTAGE

LAMP VOLTAGE

Maximum lamp wattage

Minimum lamp wattage

Lamp line

Ballast line

Maximum Lamp

Voltage

Minimum Lamp

Voltage

Operating

Point

FIGURE 22.7 Lamp and ballast characteristics.

A fraction of a second later, the glow switch is opened and the

energy stored in the ballast inductor causes a voltage spike

between the lamp electrodes (about 800 V), which ﬁnally pro-

duces the discharge breakdown. Once ignited, the lamp voltage

is lower than the line voltage and the glow switch remains

open during the normal lamp operation. Typical glow switches

are based on two bi-metal strips inside a small tube ﬁlled

with an inert gas. An external capacitor of about 10 nF is

used to enhance the glow-switch operation and also to reduce

radio interference during lamp starting. Finally, in this type

of inductive ballasts a capacitor placed across the line input is

mandatory to achieve a reasonable value of the input power

factor.

Starting voltages of high-pressure discharge lamps are nor-

mally higher than low-pressure discharge lamps and can range

from 2500 V for a lamp at room temperature to 30–40 kV to

re-ignite a hot lamp. Thus, the simple ignition system based

on the glow switch is no longer applicable for these lamps.

Figures 22.8b and c show two typical arrangements to supply

high-intensity discharge lamps. A series inductor is also used

to limit the lamp current at steady-state operation but auto-

transformers are used to attain higher voltage spikes for lamp

ignition. For higher line voltages and short distances between

starter and lamp, the inductor ballast can be used as ignition

transformer as shown in Fig. 22.8b. In other cases, a separate

igniting transformer is needed to provide higher voltage spikes

and to avoid the effect of parasitic capacitance of connection

cables (Fig. 22.8c).

The inductive ballast provides low lamp-power regula-

tion against line voltage variation and therefore it is only

recommended in those installations with low voltage ﬂuc-

tuations. When a good lamp power regulation is necessary

the circuit shown in Fig. 22.8d is normally used. This cir-

cuit is commonly named as constant wattage autotransformer

(CWA) and incorporates a capacitor in series with the lamp

to limit the discharge current. Compared with the normal